1. Field of the Invention
The present invention relates to the field of amplifiers. More specifically, the present invention relates to fast transconductance amplifiers.
2. Discussion of the Related Art
FIG. 1 is a schematic electric diagram illustrating an amplifier AMP intended to drive a capacitive load Cs, for example, the cathode of a cathode-ray tube. Generally, a resistor Rs is connected between the output of amplifier AMP and capacitor Cs.
Amplifier AMP is formed by the series-connection of an amplifier with a transconductance (G) OTA and of an amplifier-follower OP of gain unity (1). Amplifier OTA exhibits an output resistance Ro and an output capacitance Cc. The output of follower amplifier OP which forms the output of amplifier AMP is brought back into an input A of amplifier OTA via a reverse feedback resistor Rf. Input A receives an input signal via a resistor Re. Another input B of amplifier OTA is connected to a reference voltage source Vref on the order of 3 V. A voltage difference Vin is applied between the two inputs B and A via reverse feedback network Rf/Re.
Amplifier AMP provides an armature Q of capacitor Cs with a voltage variation ΔV. Variation ΔV must generally be provided with a determined slew rate Δt. For example, when load Cs is a cathode-ray screen, variation ΔV is on the order of 100 V and must occur with a relatively short time Δt of at most 10 ns. Variation ΔV on terminal Q is equal to the voltage variation at the output of amplifier OTA. To obtain a variation ΔV within a time Δt, the charge current of capacitor Cc, that is, the output current of amplifier OTA, varies from a transient or dynamic value iOUT=Cc(ΔV/ΔT). Assuming that the value of capacitor Cc is on the order of 1 pF, current iOUT must thus be equal to approximately 10 mA to obtain a variation ΔV of 100 V within a time Δt of 10 ns.
Two types of transconductance amplifiers exist. Class A amplifier, in which the quiescent current is at least equal to the transient current. Considering the preceding example, a quiescent current equal to 10 mA would then be necessary, which corresponds to a high static dissipation.
To reduce the power consumption, a class AB amplifier OTA is thus generally used, in which the common-mode current of the amplifier is small in the quiescent state and is automatically adjusted, on a variation, to the value of the transient or dynamic current.
FIG. 2 schematically illustrates a conventional embodiment of a known class AB transconductance amplifier. Amplifier OTA comprises a high-voltage stage comprising a current mirror 10 formed of two P-channel MOS transistors P1 and P2 having their sources connected to a same high supply rail Vdd, ranging between 160 and 220 V. Drain D1 of transistor P1 is connected to common gate G of transistors P1 and P2. Drain D2 of transistor P2 forms output terminal OUT of amplifier OTA. Each terminal D1 and D2 is connected to a respective output terminal C1 and C2 of an input stage 20 of amplifier OTA.
Stage 20 generally is a low-voltage stage and a cascode assembly 30 is interposed between mirror 10 and input stage 20. Cascode assembly 30 is formed of two identical N-channel MOS transistors N1 and N2 having their interconnected gates connected to the same low voltage Vcc, generally on the order of from 5 to 12 V. The drain of transistor N1 is connected to terminal D1 and the drain of transistor N2 is connected to terminal D2. The source of transistor N1 is connected to terminal C1 and the source of transistor N2 is connected to terminal C2. The structure of input stage 20 is the following.
Inputs A and B of amplifier OTA are connected to the bases of respective identical follower-assembled NPN-type bipolar transistors T1 and T2. The collectors of transistors T1 and T2 are connected to low power supply Vcc. The emitters of transistors T1 and T2 are connected to respective nodes A′ and B′.
Each of nodes A′ and B′ is connected to the base of a respective NPN-type bipolar transistor T3 and T4. Transistors T3 and T4 are identical. The collector of transistor T3 forms terminal Cl. The collector of transistor T4 forms terminal C2. A resistor (R) 22 is connected between emitters E1 and E2 of transistors T3 and T4. The respective emitters E1 and E2 are connected to the output terminal of a respective variable common-mode current source 40 and 50. Sources 40 and 50 have identical structures. Source 40 comprises, between node A′ and a reference supply rail or ground GND (0 V), a voltage source PS1, and a current source CS1. The junction point of source PS1 and CS1 is connected to the base of a bipolar PNP-type transistor T6 having its emitter connected to terminal E1 and having its collector connected to ground GND. Symmetrically, source 50 includes a voltage source PS2 and a current source CS2 series-connected between node B′ and reference rail GND. Source 50 also comprises a bipolar PNP-type transistor T7 having its base connected to the junction point of source PS2 and CS2 and having its emitter connected to node E2. Current sources CS1 and CS2 are identical. Voltage sources PS1 and PS2 are identical. Transistors T6 and T7 are identical.
The operation of the transconductance amplifier of FIG. 2 is the following. Upon occurrence of a potential difference Vin between inputs B and A of amplifier OTA, difference Vin is transmitted between nodes A′ and B′ and thus between nodes E1 and E2. Potential difference Vin between nodes E1 and E2 of resistor 22 causes the flowing of a current i, with the positive convention of the current directed from E1 to E2. The current transmitted by transistor N1 of cascode assembly 30 on terminal D1 then is I0+i, where I0 is the common-mode current set by source 40. Current I0+i is copied by mirror 10 at node D2. ΔAt the level of terminal E2, the current coming from transistor T4, and thus coming out of node D2, must be equal to 10−i, where I0 is the common-mode current set by source 50. Charge current iout of capacitor Cc is thus equal to twice current i flowing through resistor 22. The various circuit parameters are thus set according to relation iout=2i=Vin/2R=Cc (ΔV/Δt).
Value I0 of the common-mode current set by sources 40 and/or 50 automatically adapts to the value of current i. In the absence of a variation of Vin, this current is minimum, set by the characteristics of sources PS1, PS2, CS1, and CS2 of sources 40 and 50. Upon occurrence of a variation of Vin, the potential variation at nodes A′ and B′ automatically modulates the control of transistors T6 and T7 which provide the adapted current I0 for I0±i to be non-zero. Input transistors T1 and T2 are not indispensable to the circuit operation and signal Vin may be applied directly between nodes A′ and B′. However, transistors T1 and T2 enable isolating input signal Vin from identical voltage sources PS1 and PS2 connected to nodes A′ and B′.
This ability of the circuit to draw any value of common-mode current I0 necessary to the proper circuit operation causes malfunctions. For example, in the control cycle of a cathode-ray screen, it is passed through so-called blanking phases during which terminal B is maintained at a reference voltage Vref on the order of 3 volts and terminal A is directly connected to ground GND at 0 V. The 3-volt potential difference which then appears across resistor 22 with a 100-Ω value R translates as a dynamic current i on the order of 30 mA. The output current then is very high, on the order of 60 mA. The value of common-mode current 10 is automatically adjusted to be at least equal to 30 mA.
Then, transistors T6 and T7 draw the high common-mode current from the power supply, which increases the circuit power consumption. Further, in one of branches D1-C1-E1 and D2-C2-E2, for example, in branch D1-C1-E1, flows a high current I0±i on the order of 60 mA. As high a current may damage the elements of cascode assembly 30 and/or of current mirror 10, or even bipolar transistor T3.
Further, the blanking phase periodically appears on control of the screen, periodically increasing the circuit power consumption. This periodic character increases the fatigue—the wearing—and thus the damaging risks of the elements of the circuit abruptly submitted to a high current.
Moreover, the abrupt voltage variation ΔV linked to the abrupt variation of the output current to a very high value is excessive, and unnecessary.
The only known way of attempting to overcome these disadvantages is to return to a class A amplifier circuit. In such an amplifier, as compared to the class AB amplifier of FIG. 2, identical variable current sources 40 and 50 are replaced with fixed current sources. The dynamic current peaks are then impossible. This is however obtained at the cost of a high continuous power consumption.